Method and apparatus for a parallel correlator and applications thereof

ABSTRACT

A fast correlator transform (FCT) algorithm and methods and systems for implementing same, correlate an encoded data word (X 0 -X M-1 ) with encoding coefficients (C 0 -C M-1 ), wherein each of (X 0 -X M-1 ) is represented by one or more bits and each said coefficient is represented by one or more bits, wherein each coefficient has k possible states, and wherein M is greater than 1. In accordance with the invention, X 0  is multiplied by each state (C 0 ( 0 ) through C 0(k-1) ) of the coefficient C 0 , thereby generating results X 0 C 0(0)  through X 0 C 0(k-1) . This is repeating for data bits (X 1 -X M-1 ) and corresponding coefficients (C 1 -C M-1 ), respectively. The results are grouped into N groups. Members of each of the N groups are added to one another, thereby generating a first layer of correlation results. The first layer of results is grouped and the members of each group are summed with one another to generate a second layer of results. This process is repeated as necessary until a final layer of results is generated. The final layer of results includes a separate correlation output for each possible state of the complete set of coefficients (C 0 -C M-1 ). The final layer of results is compared to identify a most likely code encoded on said data word. In an embodiment, the summations are pruned to exclude summations that would result in invalid combinations of the encoding coefficients (C 0 -C M-1 ). In an embodiment, substantially the same hardware is utilized for processing in-phase and quadrature phase components of the data word (X 0 -X M-1 ). In an embodiment, the coefficients (C 0 -C M-1 ) represent real numbers. In an alternative embodiment, the coefficients (C 0 -C M-1 ) represent complex numbers. In an embodiment, the coefficients (C 0 -C M-1 ) are represented with a single bit. Alternatively, the coefficients (C 0 -C M-1 ) are represented with multiple bits (e.g., magnitude). In an embodiment, the coefficients (C 0 -C M-1 ) represent a cyclic code keying (“CCK”) code set substantially in accordance with IEEE 802.11 WLAN standard.

CROSS-REFERENCE TO RELATED APPLICATIONS

This patent application claims priority to U.S. provisional application, “Method and Apparatus for a Parallel Correlator and Applications Thereof,” Ser. No. 60/248,001, filed Nov. 14, 2000, incorporated herein by reference in its entirety.

The following application of common assignee is related to the present application, and is herein incorporated by reference in its entirety: U.S. non-provisional application, “Method and System for Down-Converting an Electromagnetic Signal, Transforms for Same, and Aperture Relationships,” Ser. No. 09/550,644, filed Apr. 14, 2000.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to matched filters and correlators and, more particularly, to a novel fast correlator transform (“FCT”) and to methods and systems for implementing same.

2. Related Art

Matched Filter Theory was introduced by D. O. North in: 1943. Others such as Van Vleck, Middleton, Weiner, and Turin, have added to the body of knowledge and application, ranging from RADAR to SONAR and communications. Although the basic theory was originally applied to continuous time systems and linear networks, the theory has been augmented to absorb discrete sampled filters and correlator operations. Indeed Paul Green pointed out in the June 1960 IRE Transactions on Information Theory, Matched Filter Issue:

“Correlation detection was studied at first as a separate subject, but the equivalence of the two operations (meaning matched filters and correlators) was soon appreciated.”

IRE Transactions on Information Theory, New York, N.Y.: Professional Group on Information, Institute of Radio Engineers, June, 1960, incorporated herein by reference in its entirety.

More recently Van Whalen and Blahut as well as others have provided proofs of mathematical equivalence of correlation or matched filtering based on a change of variables formulation.

With the surge of DSP and proliferation of VLSI CMOS circuits as well as the universal push for software radios, many discrete matched filter realizations have evolved. The most notable digital implementation is the Transversal or Finite Impulse Response Filter which may realize the time flipped impulse response of the waveform to be processed or utilized as a correlator, both which produce the equivalent result at the optimum observation instant.

A particular concern arises when multiple filtering operations are required, concurrently, as is the case for parallel correlators. The complexity, clock speeds and signal flow control typically increase cost, size, and power.

What are needed are improved methods and systems for performing matched filtering and/or correlating functions, including concurrent and/or parallel correlations.

SUMMARY OF THE INVENTION

The present invention is directed to methods and systems for performing matched filtering and/or correlating functions, including concurrent and/or parallel correlations. More particularly, the present invention is directed to a fast correlator transform (FCT) algorithm and methods and systems for implementing same. The invention is uesful for, among other things, correlating an encoded data word (X₀-X_(M-1)) with encoding coefficients (C₀-C_(M-1)), wherein each of (X₀-X_(M-1)) is represented by one or more bits and each coefficient is represented by one or more bits, wherein each coefficient has k possible states, and wherein M is greater than 1.

In accordance with the invention, X₀ is multiplied by each state (C₀₍₀₎ through C_(0(k-1))) of the coefficient C₀, thereby generating results X₀C₀₍₀₎ through X₀C_(0(k-1)). This is repeating for data bits (X₁-X_(M-1)) and corresponding coefficients (C₁-C_(M-1)), respectively. The results are grouped into N groups combinations within each of said N groups are added to one another, thereby generating a first layer of correlation results.

The first layer of results is grouped and the members of each group are summed with one another to generate a second layer of results. This process is repeated as necessary until a final layer of results is generated. The final layer of results includes a separate correlation output for each possible state of the complete set of coefficients (C₀-C_(M-1)). The results in the final layer are compared with one another to identify a most likely code encoded on said data word.

In an embodiment, the summations are pruned to exclude summations that would result in invalid combinations of the encoding coefficients (C₀-C_(M-1)). In an embodiment, substantially the same hardware is utilized for processing in-phase and quadrature phase components of the data word (X₀-X_(M-1)). In an embodiment, the coefficients (C₀-C_(M-1)) represent real numbers. In an alternative embodiment, the coefficients (C₀-C_(M-1)) represent complex numbers. In an embodiment, the coefficients (C₀-C_(M-1)) are represented with a single bit. Alternatively, the coefficients (C₀-C_(M-1)) are represented with multiple bits (e.g., magnitude). In an embodiment, the coefficients (C₀-C_(M-1)) represent a cyclic code keying (“CCK”) code set substantially in accordance with IEEE 802.11 WLAN standard.

Further features and advantages of the invention, as well as the structure and operation of various embodiments of the invention, are described in detail below with reference to the accompanying drawings. It is noted that the invention is not limited to the specific embodiments described herein. Such embodiments are presented herein for illustrative purposes only. Additional embodiments will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. The drawing in which an element first appears is typically indicated by the leftmost digit(s) in the corresponding reference number.

BRIEF DESCRIPTION OF THE DRAWINGS/FIGURES

The present invention will be described with reference to the accompanying drawings wherein:

FIG. 1 is a block diagram of an example discrete transversal matched filter or correlator, in which the present invention can be implemented;

FIG. 2 is an expanded view of the summation function illustrated in FIG. 1;

FIG. 3A illustrates example correlation kernels, in accordance with an aspect of the present invention;

FIG. 3B illustrates example first and second layers of an FCT processing hierarchy in accordance with an aspect of the invention;

FIG. 3C illustrates an example signal flow diagram for the first and second layers illustrated in FIG. 3B;

FIG. 4A illustrates a relationship between a conventional parallel correlator approach and Equation 7;

FIG. 4B illustrates an example matrix form of coefficients for a parallel correlator in accordance with an aspect of the present invention;

FIG. 5 illustrates an example complex fast Hadamard Transform;

FIG. 6 illustrates an example parallel magnitude compare operation in accordance with an aspect of the invention;

FIG. 7 illustrates an example final layer of an FCT processing hierarchy in accordance with an aspect of the invention;

FIG. 8 illustrates an example process flowchart for implementing an FCT algorithm in accordance with an aspect of the present invention.

DETAILED DESCRIPTION OF THE INVENTION

TABLE OF CONTENTS I. Introduction II. Example Environment: 802.11 III. Fast Correlator Transform and Correlator Kernels IV. Mathematical Formulation V. Comparisons to the Hadamard Transform VI. Maximum Likelihood Decoding (AWGN, no Multipath) A. Magnitude Comparator VII. Example Methods for Implementing the FCT Alrgorithm VIII. CCK Chip Code Words IX Conclusion I. Introduction

FIG. 1 is a block diagram of an example discrete transversal matched filter or correlator 100, also referred to herein as a finite impulse response (“FIR”) filter 100. The FIR filter 100 includes a set of multipliers 104 that multiply data words X_(i) by coefficients C. The FIR filter 100 also a final accumulate or summation function 102. The matched filter 100 implements the following discrete sampling equation; $\begin{matrix} {y_{i} = {\sum\limits_{k = 0}^{n - 1}\quad{c_{k}x_{({i - k})}}}} & \left( {{Eq}.\quad 1} \right) \end{matrix}$

In Eq. 1, X_(i) are typically derived from a sampling device such as an A/D converter and can be represented as signed magnitude, two's complement, or other binary representation in parallel bit format. Likewise the multiplier tap coefficients C₀ . . . C_(n) can be 1 bit values or k bit (soft) values depending on the application. In the case of soft value implementations, the multiplier functions can represent considerable hardware or algorithmic complexity. The final summation can be implemented using discrete adder technologies or floating point processors. The entire architecture can be implemented in hardware, software, or a hybrid of each.

A particular concern arises when multiple filtering operations are required, concurrently, as is the case for parallel correlators. The complexity, clock speeds and signal flow control typically increase cost, size, and power. Hence, efficient architectures are often pursued to streamline the implementation and thereby differentiate product offerings.

The present invention is directed to a novel fast correlator transform (“FCT”) algorithm that reduces the number of additions for parallel correlation, as compared to conventional techniques. The present invention is also directed to methods and systems for implementing the novel FCT algorithm. Conventional techniques are typically of less universal application and restricted to certain signaling schemes. The present invention, on the other hand, is applicable to a variety of parallel matched filter and/or correlator operations.

The present invention is as efficient or more efficient than the Fast Walsh Transform, currently applied as the “state of the art,” and is more universally applicable to signaling schemes employing higher order signaling spaces such as MQAM, CDMA, etc. In addition, classical multi-path processor algorithms are more easily applied using the classical correlator/matched filter kernel, in accordance with the present invention, rather than the conventional modified Fast Walsh Transform.

II. Example Environment: 802.11

The present invention is described herein as implemented in an example environment of an IEEE 802.11b 11 MBPS physical layer signaling scheme. IEEE 802.11 is a well-known communications standard and is described in, for example, “Medium Access Control (MAC) and Physical (PHY) Specifications,” ANS/IEE Std 802.11, published by IEEE, (1999 Ed.), and incorporated herein by reference in its entirety.

The present invention is not, however, limited to the IEEE 802.11 communications standard. Based on the description herein, one skilled in the relevant art(s) will understand that the present invention can be implemented for a variety of other applications as well. Such other applications are within the spirit and scope of the present invention.

The 802.11 signaling scheme of interest is based on Cyclic Code Keying (“CCK”) derived from Walsh/Hadamard functions. A restricted set within the available coding space was selected so that the Fast Walsh Transform could be utilized to implement an efficient correlator architecture. Originally, both Harris and Lucent could not figure out how to apply a classical parallel matched filter or correlator, efficiently, to process the waveforms. The current coding space provides for 64 code words. Harris erroneously predicted that a classical parallel matched filter approach would require 8×64=512 complex additions since each code word is 8 bits, on 1 and Q and there are 64 code words. However, the true estimate is 7×64=448 complex additions as illustrated in the example 8-way adder tree illustrated in FIG. 2.

FIG. 2 is an expanded view of the final accumulate or summation function 102 in FIG. 1, part of the FIR filter 100. Notice that only 7 adders are required for the example implementation. Nevertheless, 448 complex additions represent a significant number of operations. Lucent, Harris/Intersil, and Alantro apply the Fast Walsh Transform (“FWT”) to the CCK code set to reduce the correlation operation down to 112 complex multiplies due to the restriction placed on the code set.

The FWT is actually more of a decoder than a correlator. It reduces to the performance of a correlator for this specific application if the coefficients in the butterfly branches are weighted with simple hard coded values, i.e., 1, −1, j, −j. The imaginary numbers are included for the case of complex signaling.

The FCT algorithm, according to the present invention, is truly a correlation or matched filter operation and can be applied with soft value weighting or hard value weighting. Furthermore, the present invention is intuitively satisfying, possessing direct correspondence of matched filter tap weights or coefficients maintained throughout the hierarchical structure. This permits easy extension of the matched filter concept to accommodate channel equalization, MLSE, and other adaptive applications.

III. Fast Correlator Transform and Correlator Kernels

The available coding space for a 16 bit word is 2¹⁶=65536. CCK whittles this space down to a code set formed by 8 in-phase (“I”) chip and 8 quadrature phase (“Q”) chip complex symbols. 64 code words are retained, carrying 6 bits of information. 2 more bits are contained in the complex QPSK representation, for a total of 8 bits of information.

Suppose then that 8 samples of an input vector X₀, X₁, . . . X₇ are associated with optimal sampling instants from the output of a chip matched filter. Each of these samples would normally be weighted by the coefficients C₀ . . . C₇ then assimilated as illustrated in FIGS. 1 and 2.

In the 802.11 example, C₀ . . . . C₇ correspond to samples of the CCK code set symbols. The unknowns X₀ . . . X₇ are noisy input samples from which a specific code must be extracted. Conceptually, 64 such complex filtering operations should be performed concurrently since a specific code word cannot be predicted a priori. The largest output from the parallel bank of correlators would then be selected as the most likely code word correlation match at that particularly symbol sample time.

In accordance with the present invention, a general philosophy for correlation is imposed for partitioned segments of code words. In an embodiment, the code word correlations are divided into sub-sets. In the illustrated example embodiment, the code word correlations are divided into sample pairs. The present invention is not, however, limited to this example embodiment. In other embodiments, the code word correlations are divided into triplets, quintuplets, and/or any other suitable sub-sets.

Combinations of the code word correlation sub-sets are then provided to correlation kernels. FIG. 3A illustrates example correlation kernels 302 a-302 d, in accordance with an aspect of the present invention.

The correlation kernels 302 a-302 d represent all or substantially all possible correlation options for the first 2 samples X₀, X₁. In a similar fashion the remaining groupings of double samples (X₂, X₃), (X₄, X₅), (X₆, X₇) will also spawn their own set of 4 correlation kernels.

The number of separate correlation kernels spawned is determined by the number of correlated samples per summation, the number of correlation variations possible, and, in an embodiment, the number of invalid combinations of correlated samples. In the present example, each coefficient has two possible states (i.e., hard value weighting). Thus each subset correlation generates 4 outputs. In alternative embodiments, soft weighting is implemented, where each coefficient is represented with multiple bits (e.g., magnitude representation).

In an embodiment, binary antipodal signaling in accordance with the present invention is implemented in accordance with Eq. 2. $\begin{matrix} {N_{k} = {\frac{n!}{{r!}{\left( {n - r} \right)!}} - L}} & \left( {{Eq}.\quad 2} \right) \end{matrix}$

The result for the example environment described above, using 2 input summers, is shown in Eq. 3: $\begin{matrix} {N_{k} = {\frac{4!}{{2!}{(2)!}} - 2}} & \left( {{Eq}.\quad 3} \right) \end{matrix}$

-   -   wherein:     -   n is the number of uniquely available summer inputs;     -   r is the number of summing inputs per kernel; and

-   L is the number of invalid combinations.

N_(k) is thus the number of correlation kernels and therefore the number of adders or summers required. Groupings of two correlation samples provide for convenient binary expansion. As stated above, however, the present invention is not limited to groupings of two.

The term L represents the number of invalid or disallowed combinations. For instance, X₀C₀ and —X₀C₀ is an invalid combination when added and therefore can be subtracted from the total number of combinations. In an embodiment, 3 way adders or summers are utilized. In other embodiments, other algorithmic partitioning is utilized. For the current example, partitioning in powers of 2 is convenient and attractive in terms of potential hardware implementation.

FIG. 3B illustrates example first and second layers 304 and 306, respectively, of an FCT processing hierarchy in accordance with an aspect of the invention.

The second layer 306 includes 32 additions 308, of which 308 a through 308 p are illustrated, to process combinations of correlations from the first layer 304 of correlation kernels. The first layer 304 includes 16 additions related to the first 4 sample correlations X₀C₀ . . . X₃C₃, and 16 additions related to the 2^(nd) 4 sample correlations. Hence, the first layer 304 of kernels includes 16 adders 302 and the second layer possesses 32 adders 308. Once the second layer 306 is accomplished, each term that results includes 4 correlation components.

Note that 4 unique samples X₀ . . . X₃ at the system input spawns 24 unique 4-tuple correlations at the second layer 306 of kernel processing. The corresponding number of additions is calculated for the 4 sample correlation sequences from Eq. 4: $\begin{matrix} {N = {{\frac{8!}{{2!}{6!}} - {2\left( \frac{4!}{{2!}{2!}} \right)}} = 16}} & \left( {{Eq}.\quad 4} \right) \end{matrix}$

At this point it is important to realize that all that is required is one more level of processing to obtain correlation terms consisting of 8 components which represents a full length correlation. However, it must also be recognized that there are 16 upper 4-tuple correlations as well as 16 lower 4-tuple correlations, which if exploited for all combinations in this case would yield 256 addition operations! Fortunately the CCK code set is well defined and possesses only 64 valid 8 chip component correlations. Hence, the final layer, illustrated in FIG. 7, is pruned to perform only 64 unique addition operations. Thus, a total (upper bound) number of adders used for the algorithm are: 16 (first hierarchical layer)+32 (second layer)+64 (third layer)=112

This is a remarkable result because a conventional parallel matched filter or correlator bank would require 448 complex additions. Theoretically, 112 is the upper bound. However, in practice, the Trellis may be pruned to a maximum of 78 additions on the I and 78 and the Q.

FIG. 3C illustrates an example signal flow diagram for the FCT algorithm through the first 2 layers 304 and 306. In accordance with the example above, there are 8 input samples and 32 output correlation options through the first 2 layers 304 and 306. Correlation combinations from the upper and lower 16 4-tuples provide a final trellis with 64 8-tuple options, each representing a different CCK symbol. The option having the highest output is selected during each symbol cycle as the most likely CCK symbol. In the example embodiment, the correlation operation utilizes I and Q matched filters since the symbols are complex.

IV. Mathematical Formulation

In an embodiment, a receiver in accordance with the present invention receives a waveform and an analog-to-digital converter function samples the received waveform and converts the waveform down to baseband. The received sampled waveform may be represented from Eq. 5: X _(i) =S _(i) +N _(i)  (Eq. 5)

Where S_(i) represents samples from the received signal and N_(i) represent noise sampled from an average white Gausian noise (“AWGN”) process. This equation does not account for multi-path. The samples can be considered as complex, with I and Q components. The receiver output can be represented by Eq. 6: $\begin{matrix} {Y_{i} = {\sum\limits_{k = 0}^{n - 1}\quad{C_{k}{X\left( {i - k} \right)}}}} & \left( {{Eq}.\quad 6} \right) \end{matrix}$

The coefficients, C_(k), are considered constant for the nominal case of an AWGN channel. n is the FIR filter or correlator depth. For a case of m correlators operating on X_(i) in parallel, Eq. 6 becomes Eq. 7: $\begin{matrix} {Y_{i,{m - 1}} = {\sum\limits_{k = 0}^{n - 1}\quad{C_{k,{m - 1}}{X\left( {i - k} \right)}}}} & \left( {{Eq}.\quad 7} \right) \end{matrix}$

The mth correlator branch then contains correlator coefficients uniquely related to that branch.

FIG. 4A illustrates a conventional parallel correlator approach and relates it to Eq. 7. The present invention breaks the sum over n−1 into smaller sums, typically, although not necessarily, sums of 2. The present invention applies all, or substantially all potential cross correlations and carries the 4 results forward to a subsequent level of processing. An example mathematical formulation of this operation is provided in Eq. 8; $\begin{matrix} {{{Y(i)}_{l,v,p,u} = {{\sum\limits_{k = 0}^{1}\quad{C_{k,l}{X\left( {i - k} \right)}}} + {\sum\limits_{k = 2}^{3}\quad{C_{k,v}{X\left( {i - k} \right)}}} + {\sum\limits_{k = 4}^{5}\quad{C_{k,p}{X\left( {i - k} \right)}}} + {\sum\limits_{k = 6}^{7}\quad{C_{k,u}{X\left( {i - k} \right)}}}}}\quad} & \left( {{Eq}.\quad 8} \right) \end{matrix}$

Where l, v, p, and u may be considered as indices to select different coefficients. All indices should be considered in terms of a final valid code word correlation. In the 802.11 case, 256 correlation sequences are defined by Eq. 8, but the options are sifted to correspond only to valid CCK code words. FIG. 4B illustrates an example matrix form of coefficients for a parallel correlator according to the present invention.

The coefficients all take on the values of +/−1 for the examples herein. The indices are permitted to contain zeros for consistency with the original FIR formulation. The FCT sub-matrices however are simply; $\begin{matrix} {C_{k,l} = {C_{k,v} = {C_{k,p} = {C_{k,u} = \begin{bmatrix} 1 & {- 1} \\ {- 1} & 1 \end{bmatrix}}}}} & \left( {{Eq}.\quad 9} \right) \end{matrix}$

The indices l, v, p, u are manipulated specifically to match the coefficients to the desired code words at the Y(i)_(l,v,p,u) outputs. The indices also uniquely specify the trajectory through the signal flow path. Breaking up the original parallel matched filter coefficient matrix into the smaller 2×2 matrix permits the FCT algorithm to exploit redundant correlation operations.

V. Comparisons to the Hadamard Transform

An FCT algorithm trellis, in accordance with the present invention, is described above. A corresponding length 4 complex fast Hadamard Transform is illustrated in FIG. 5. As with the FCT algorithm, there are two such trellis structures corresponding to 8 input samples and 32 outputs. The 32 outputs include two 16 wide 4-tuple groupings, which are then utilized in combinations to produce the final 54 8-tuple correlations.

There are distinct differences between the two algorithms. For example, the FCT algorithm can function on arbitrary correlation words, even in matched filter scenarios, while the FWT requires certain signal coding structure. Also, the FCT algorithm permits greater efficiency because the number of adds may be tailored for a specific code set or application.

Harris and Lucent advertise an efficiency of 112 complex additions for CCK demodulation, which amounts to 2×112 additions. The bounding case for 64 arbitrary correlations with the FCT was shown to be 112, maximum. It turns out that the specific case of CCK may be accommodated using a pruned FCT algorithm having 78 additions on the in-phase correlation band and 78 additions on the quadrature correlation bank. Therefore, the FCT possesses greater redundancy for the CCK application, which is exploited to implement significantly more efficient structures.

VI. Maximum Likelihood Decoding (AWGN, no Multipath)

The modified Fast Walsh/Hadamard Transform implements a complex trellis decoder for which maximum scores may be compared at the trellis output. Thus there are 64 distinct outputs of the trellis which are preferred based on Euclidean distance calculations along the trellis trajectories. Only certain specific trajectories are considered up through the second tier of the trellis. The distance properties for the decoding trellis are also Euclidean for the in-phase and quadrature phase correlators. However, it is important to realize that the total distance should be calculated within the complex plane rather than simply on I independently of Q. That is, scoring is based on Eq. 10. Distance≡{square root}{square root over (I _(score) ² +Q _(score) ²)}  (Eq. 10)

This comes from the fact that there are pairs of I and Q chip code words which are dependent. That is the nature of the complex Walsh-Hadamard codes. Fortunately, a sufficient statistic exists which does not require the square root operation. Simply calculating the sum of the squares or estimating the vector magnitude will suffice. In this manner then the total distance or weighting through complex space is calculated. The largest output out of the 64 complex operations (weighting scenario) then becomes the most likely 8 chip complex code.

A. Magnitude Comparator

In order to determine which code symbol was most likely encoded, a magnitude compare operation is performed on the outputs from the summer 102 (FIG. 1). A variety of types of magnitude compare operations can be performed.

FIG. 6 illustrates an example parallel magnitude compare operation in accordance with an aspect of the invention. In operation, the I and Q inputs, 8 bits wide each, for example, are squared and summed at the correlator outputs to form 64 scores. These 64 scores are compared and the largest result selected as the maximum likelihood symbol estimate. Each of the 64 outputs are assigned their corresponding chip code-to-6-bit data map. An additional di-bit is decoded from the differential phase decoder. In an embodiment, the squaring operation results in 16 bit output value when the inputs from each I and Q correlator are truncated to an extent reasonable to minimize the compare tree. In an embodiment, a parallel compare tree utilizes log₂ (64)−1 compares to obtain the most likely result.

In an embodiment, the magnitude compare operation illustrated in FIG. 6 utilizes a flag at each level of compare to indicate the winning local score at that level. The winning local score can be traced from the output back to one of the 64 original input correlation scores to decide which 6-bit word is most likely. In an embodiment, outcomes of tie scores at one or more levels are arbitrarily determined. In an embodiment, magnitude compare operations are performed with an adder/subtractor to create the result C=A−B, where A and B are inputs.

Another magnitude compare technique that can be utilized is referred to herein as space slicing, which includes the steps of examining the MSB of the correlator outputs, and throwing away results not active in the MSB. If none are active in the MSB then the next MSB is compared, so on and so forth. Any surviving correlator outputs are compared in the next most significant MSB in succession until all necessary compares are exhausted. This technique is useful because it requires only 1-bit compares at each level down to the final compare. In an embodiment, 1 bit compares are performed with an exclusive OR gate. Generally, there is no deterministic way to predict the number of surviving compares which may be passed on to the next level, but the maximum number typically reduces by a factor of 2 at each level. This approach relies on a statistical distribution of scores, which may permit rapid sifting. If all of the distances are similar in magnitude then sifting typically requires more operations. For instance, if all 64 distance calculations/scores possess an active MSB then the first round of sifting will not eliminate any scores and all scores are then be compared in the next MSB. Although this is not likely to occur, it should be anticipated for associated hardware realization.

VII. Example Methods for Implementing the FCT Alrgorithm

FIG. 8 illustrates an example process flowchart 800 for implementing the FCT algorithm in accordance with an aspect of the present invention. For illustrative purposes, the flowchart 800 is described herein with reference to one or more of the drawing figures described above. The invention is not, however, limited to the examples illustrated in the drawings. Based on the description herein, one skilled in the relevant art(s) will understand that the invention can be implemented in a variety of ways.

The example process flowchart 800 illustrates a method for correlating an encoded data word (X₀-X_(M-1)) with encoding coefficients (C₀-C_(M-1)), wherein each of (X₀-X_(M-1)) is represented by one or more bits and each said coefficient is represented by one or more bits, wherein each coefficient has k possible states, wherein M is greater than 1.

The process begins with step 802, which includes multiplying X₀ with each state (C₀₍₀₎ through C_(0(k-1))) of the coefficient C₀, thereby generating results X₀C₀₍₀₎ through X₀C_(0(k-1)). This is illustrated, for example, in FIGS. 3A, 3B, and 3C just prior to the kernels 302A, B, C, and D.

Step 804 includes repeating step 802 for data bits (X₁-X_(M-1)) and corresponding coefficients (C₁-C_(M-1)), respectively. This is illustrated, for example, in FIGS. 3B, and 3C just prior to the kernels 302E through 302Q.

Step 806 includes grouping the results of steps 802 and 804 into N groups and summing combinations within each of said N groups, thereby generating a first layer of correlation results. This is illustrated, for example, in FIGS. 3A, 3B, and 3C by the kernels 302, and the resultant first layer of results 307.

Step 808 includes grouping the results of step 806 and summing combinations of results within each group to generate one or more additional layers of results, and repeating this process until a final layer of results includes a separate correlation output for each possible state of the complete set of coefficients (C₀-C_(M-1)). This is illustrated in FIG. 3C and FIG. 7, where the summers 306 generate a second layer 310, the FCT final output trellis 702 (FIG. 7) provides separate outputs for each possible state of the complete set of coefficients (C₀-C_(M-1)) in a final layer 704.

In an embodiment, steps 806 and 808 include the step of omitting summations that would result in invalid combinations of the encoding coefficients (C₀-C_(M-1)). This is illustrated, for example, in FIG. 7, wherein the second level of results 310 omits the following combinations:

-   C₄C₅C₆(—C₇); -   C₄C₅(—C₆)(—C₇); -   (—C₄)C₅C₆(—C₇); -   (—C₄)C₅(—C₆)(—C₇); -   C₄(—C₅)C₆(—C₇); -   C₄(—C₅)(—C₆)(—C₇); -   (—C₄)(—C₅)C₆(—C₇); and -   (—C₄)(—C₅)(—C₆)(—C₇).

In this example, the omissions eliminate performing summations for combinations that are invalid in light of the CCK code. In other embodiments, different combinations may or may not be omitted based on particular codes.

Step 810 includes comparing magnitudes of said separate correlation outputs, thereby identifying a most likely code encoded on said data word. This is illustrated, for example, in FIG. 6, by the example parallel magnitude compare operation.

In an embodiment, the process flowchart 800 further includes the step of performing steps (1) through (5) using substantially the same hardware for in-phase and quadrature phase components of the data word (X₀-X_(M-1)).

In an embodiment, the coefficients (C₀-C_(M-1)) represent real numbers.

In an alternative embodiment, the coefficients (C₀-C_(M-1)) represent complex numbers.

In an embodiment, the coefficients (C₀-C_(M-1)) are represented with a single bit. Alternatively, the coefficients (C₀-C_(M-1)) are represented with multiple bits (e.g., magnitude).

In an embodiment, the coefficients (C₀-C_(M-1)) represent a cyclic code keying (“CCK”) code set substantially in accordance with IEEE 802.11 WLAN standard, illustrated in the tables below.

In an embodiment, as illustrated in one or more of the prior drawing figures, M equals 8, N equal 4, and the coefficients (C₀-C_(M-1)) have two states, plus and minus.

VIII. CCK Chip Code Words

Tables are provided below that illustrate source-input data symbols, 8-bits long (d₀-d₇), and corresponding in-phase and quadrature phase 8 chip symbols used for CCK. The complex chip notation is provided for reference. In addition the algorithm flow diagram 4-tuple sums are provided since the last level of flow diagram becomes complex and difficult to follow. B₀ . . . B₃₁ are the 4-tuple intermediate correlation results relating to the signal flow diagrams presented for the correlator algorithm. Two branch 4-tuples form the final output result for each correlator. B₀ . . . B₁₅ provide options for the first branch component to form a final output correlator 8-tuple while B₁₆ . . . B₃₁ provide the second branch component or second 4-tuple. For instance, Table 1 illustrates an example build-up: TABLE 1 4-tuple Designator 4-tuple Coefficient Sequence B₆ −1, 1, 1, −1 B₂₈ 1, 1, −1, −1 4-tuple Combination Final 8-tuple Correlator Coefficient Sequence B₆ + B₂₈

−1, 1, 1, −1, 1, 1, −1, −1

Logical zeros become weighted by an arithmetic value, −1. In this manner the optimum correlator trajectory for a particular chip sequence is projected through the correlator trellis. The example above corresponds to the in-phase correlator waiting for an originally transmitted data sequence d₀ . . . d₇ of 0,0,1,0,1,0,1,0. For this example, that particular branch represents a correlation provided in Eq. 11; y ₄₂ =x ₀(−1)+x ₁(1)+x ₂(1)+x ₃(−1)+x ₄(1)+x ₅(1)+x ₆(−1)+x ₇(−1)  (Eq. 11)

x₀ . . . x₇ represent corrupted or noisy input signal samples. When the x₁ match the coefficients significantly then that 8-tuple (1 of 64) output possesses maximum weighting and is declared most likely by the magnitude comparator. Another strategy seeks to minimize the distance between the x₁ and c₁. The process is similar in either case.

Table 2 illustrates example in-phase and quadrature 4-tuple combinations. It is noted that the examples provided in Tables 1 and 2 are provided for illustrative purposes and are not limiting. Other specific examples will be apparent to persons skilled in the relevant arts based on the teachings herein, and such other examples are within the scope and spirit of the invention. TABLE 2 d0 d1 d2 d3 4-tuple 4-tuple d4 d5 d6 d7 In phase Combination Quadrature Combination Complex D₀ 00000000 11101101  B₂ + B₂₀ 11101101  B₂ + B₂₀ 111−111−11 D₁ 00000001 00011101 B₁₃ + B₂₀ 11101101  B₂ + B₂₀ jjj−j11−11 D₂ 00000010 00011101 B₁₃ + B₂₀ 00011101 B₁₃ + B₂₀ −1−1−1111−11 D₃ 00000011 11101101  B₂ + B₂₀ 00011101 B₁₃ + B₂₀ j−j−jj11−11 D₄ 00000100 00100001 B₁₄ + B₂₃ 11101101  B₂ + B₂₀ jj1−1jj−11 D₅ 00000101 00010001 B₁₃ + B₂₃ 00101101 B₁₄ + B₂₀ −1−1j−jjj−11 D₆ 00000110 11010001  B₁ + B₂₃ 00011101 B₁₃ + B₂₀ −j−j−11jj−11 D₇ 00000111 11100001  B₂ + B₂₃ 11011101  B₁ + B₂₀ 11−jjjj−11 D₈ 00001000 00100001 B₁₄ + B₂₃ 00100001 B₁₄ + B₂₃ −1−11−1−1−1−11 D₉ 00001001 11010001  B₁ + B₂₃ 00100001 B₁₄ + B₂₃ −j−jj−j−1−1−11 D₁₀ 00001010 11010001  B₁ + B₂₃ 11010001  B₁ + B₂₃ 11−11−1−1−11 D₁₁ 00001011 00100001 B₁₄ + B₂₃ 11010001  B₁ + B₂₃ jj−jj−1−1−11 D₁₂ 00001100 11101101  B₂ + B₂₀ 00100001 B₁₄ + B₂₃ −j−j1−1−j−j−11 D₁₃ 00001101 11011101  B₁ + B₂₀ 11100001  B₂ + B₂₃ 11j−j−j−j−11 D₁₄ 00001110 00011101 B₁₃ + B₂₀ 11010001  B₁ + B₂₃ jj−11−j−j−11 D₁₅ 00001111 00101101 B₁₄ + B₂₀ 00010001 B₁₃ + B₂₃ −1−1−jj−j−j−11 D₁₆ 00010000 01000111  B₇ + B₁₇ 11101101  B₂ + B₂₀ j1j−1j1−j1 D₁₇ 00010001 00010111 B₁₃ + B₁₇ 01001101  B₇ + B₂₀ −1j−1−jj1−j1 D₁₈ 00010010 10110111  B₈ + B₁₇ 00011101 B₁₃ + B₂₀ −j−1−j1j1−j1 D₁₉ 00010011 11100111  B₂ + B₁₇ 10111101  B₈ + B₂₀ 1−j1jj1−j1 D₂₀ 00010100 00000011 B₁₅ + B₁₉ 01100101  B₆ + B₂₁ −1jj−1−1j−j1 D₂₁ 00010101 10010011  B₉ + B₁₉ 00000101 B₁₅ + B₂₁ −j−1−1−j−1j−j1 D₂₂ 00010110 11110011  B₀ + B₁₉ 10010101  B₉ + B₂₁ 1−j−j1−1j−j1 D₂₃ 00010111 01100011  B₆ + B₁₉ 11110101  B₀ + B₂₁ j11j−1j−j1 D₂₄ 00011000 10001011 B₁₁ + B₁₈ 00100001 B₁₄ + B₂₃ −j−1j−1−j−1−j1 D₂₅ 00011001 11011011  B₁ + B₁₈ 10000001 B₁₁ + B₂₃ 1−j−1−j−j−1−j1 D₂₆ 00011010 01111011  B₄ + B₁₈ 11010001  B₁ + B₂₃ j1−j1−j−1−j1 D₂₇ 00011011 00101011 B₁₄ + B₁₈ 01110001  B₄ + B₂₃ −1j1j−j−1−j1 D₂₈ 00011100 11001111  B₃ + B₁₆ 10101001 B₁₀ + B₂₂ 1−jj−11−j−j1 D₂₉ 00011101 01011111  B₅ + B₁₆ 11001001  B₃ + B₂₂ j1−1−j1−j−j1 D₃₀ 00011110 00111111 B₁₂ + B₁₆ 01011001  B₅ + B₂₂ −1j−j11−j−j1 D₃₁ 00011111 10101111 B₁₀ + B₁₆ 00111001 B₁₂ + B₂₂ −j−11j1−j−j1 D₃₂ 00100000 01000111  B₇ + B₁₇ 01000111  B₇ + B₁₇ −11−1−1−1111 D₃₃ 00100001 10110111  B₈ + B₁₇ 01000111  B₇ + B₁₇ −jj−j−j−1111 D₃₄ 00100010 10110111  B₈ + B₁₇ 10110111  B₈ + B₁₇ 1−111−1111 D₃₅ 00100011 01000111  B₇ + B₁₇ 10110111  B₈ + B₁₇ j−jjj−1111 D₃₆ 00100100 10001011 B_(11 + B) ₁₈ 01000111  B₇ + B₁₇ −jj−1−1−jj11 D₃₇ 00100101 10111011  B₈ + B₁₈ 10000111 B₁₁ + B₁₇ 1−1−j−j−jj11 D₃₈ 00100110 01111011  B₄ + B₁₈ 10110111  B₈ + B₁₇ j−j11−jj11 D₃₉ 00100111 01001011  B₇ + B₁₈ 01110111  B₄ + B₁₇ −11jj−jj11 D₄₀ 00101000 10001011 B₁₁ + B₁₈ 10001011 B₁₁ + B₁₈ 1−1−1−11−111 D₄₁ 00101001 01111011  B₄ + B₁₈ 10001011 B₁₁ + B₁₈ j−j−j−j1−111 D₄₂ 00101010 01111011  B₄ + B₁₈ 01111011  B₄ + B₁₈ −11111−111 D₄₃ 00101011 10001011 B₁₁ + B₁₈ 01111011  B₄ + B₁₈ −jjjj1−111 D₄₄ 00101100 01000111  B₇ + B₁₇ 10001011 B₁₁ + B₁₈ j−j−1−1j−j11 D₄₅ 00101101 01110111  B₄ + B₁₇ 01001011  B₇ + B₁₈ −11−j−jj−j11 D₄₆ 00101110 10110111  B₈ + B₁₇ 01111011  B₄ + B₁₈ −jj11j−j11 D₄₇ 00101111 10000111 B₁₁ + B₁₇ 10111011  B₈ + B₁₈ 1−1jjj−j11 D₄₈ 00110000 11101101  B₂ + B₂₀ 01000111  B₇ + B₁₇ −j1−j−1−j1j1 D₄₉ 00110001 10111101  B₈ + B₂₀ 11100111  B₂ + B₁₇ 1j1−j−j−j1 D₅₀ 00110010 00011101  B₃ + B₂₀ 10110111  B₈ + B₁₇ j−1j1−j1j1 D₅₁ 00110011 01001101  B₇ + B₂₀ 00010111  B₃ + B₁₇ −1−j−1j−j1j1 D₅₂ 00110100 10101001 B₁₀ + B₂₂ 11001111  B₃ + B₁₆ 1j−j−11jj1 D₅₃ 00110101 00111001 B₁₂ + B₂₂ 10101111 B₁₀ + B₁₆ j−11−j1jj1 D₅₄ 00110101 01011001  B₅ + B₂₂ 00111111 B₁₂ + B₁₆ −1−jj11jj1 D₅₅ 00110111 11001001  B₃ + B₂₂ 01011111  B₅ + B₁₆ −j1−1j1jj1 D₅₆ 00111000 00100001 B₁₄ + B₂₃ 10001011 B₁₁ + B₁₈ j−1−j−1j−1j1 D₅₇ 00111001 01110001  B₄ + B₂₃ 00101011 B₁₄ + B₁₈ −1−j1−jj−1j1 D₅₈ 00111010 11010001  B₁ + B₂₃ 01111011  B₄ + B₁₈ −j1j1j−1j1 D₅₉ 00111011 10000001 B₁₁ + B₂₃ 11011011  B₁ + B₁₈ 1j−1jj−1j1 D₆₀ 00111100 01100101  B₆ + B₂₁ 00000011 B₁₅ + B₁₉ −1−j−j−1−1−jj1 D₆₁ 00111101 11110101  B₀ + B₂₁ 01100011  B₆ + B₁₉ −j11−j−1−jj1 D₆₂ 00111110 10010101  B₉ + B₂₁ 11110011  B₀ + B₁₉ 1jj1−1−jj1 D₆₃ 00111111 00000101 B₁₅ + B₂₁ 10010011  B₉ + B₁₉ j−1−1j−1−jj1 IX Conclusion

While various embodiments of the present invention have been described above, it should be understood that they have been presented by way of example only, and not limitation. Other embodiments are possible and are covered by the invention. 

1-32. (canceled)
 33. A wireless local area network (WLAN) device, comprising: means for generating in parallel the correlation of a data word vector (X₀, . . . , X_(M-1)) with each valid combination of a coefficient set (C₀, . . . , C_(M-1)); and means for identifying the correlation corresponding to a code encoded on the data word.
 34. The WLAN device of claim 33, wherein the means for generating in parallel the correlation of a data word vector (X₀, . . . , X_(M-1)) with each valid combination of a coefficient set (C₀, . . . , C_(M-1)) includes: means for multiplying each element of the data word vector with each state of a corresponding coefficient in the coefficient set (C₀, . . . , C_(M-1)) to generate a plurality of outputs, wherein each coefficient in the coefficient set has one or more possible states; means for summing a plurality of groups of multiplying means outputs to produce a set of first correlation results in parallel, wherein each group of multiplying means outputs includes a different combination of multiplying means outputs; means for summing a plurality of groups of first correlation results to produce a set of second correlation results in parallel, wherein each group of first correlation results includes a different combination of first correlation results; and means for summing a plurality of groups of second correction results to produce in parallel the correlation of a data word vector (X₀, . . . , X_(M-1)) with each valid combination of the coefficient set (C₀, . . . , C_(M-1)).
 35. The WLAN device of claim 33, wherein the means for identifying the correlation corresponding to a code encoded on the data word includes: means for comparing magnitudes of the correlations to select the maximum result.
 36. The WLAN device of claim 33, wherein the means for summing a plurality of groups of multiplying means outputs, includes: means for identifying valid combinations of multiplying means outputs; and means for forming groups having valid combinations of multiplying means outputs.
 37. The WLAN device of claim 33, wherein the means for summing a plurality of groups of first correlation results, includes: means for identifying valid combinations of first correlation results; and means for forming groups having valid combinations of first correlation results.
 38. The WLAN device of claim 33, wherein the means for summing a plurality of groups of second correction results, includes: means for identifying valid combinations of second correlation results; and means for forming groups having valid combinations of second correlation results.
 39. A wireless local area network (WLAN) device, comprising: a plurality of inputs, one for each element of an encoded data word vector (X₀, . . . X_(M-1)); a multiplier module having a coefficient set (C₀, . . . , C_(M-1)), wherein each coefficient in the coefficient set has one or more possible states, wherein the multiplier module receives the plurality of inputs and multiplies each input with each state of a corresponding coefficient; a first summation module including a plurality of summers, each summer coupled to a different group of multiplier module outputs, wherein each group of multiplier module outputs represents a valid combination of multiplier module outputs, and wherein a first set of correlation results are output in parallel from the first summation tier; one or more additional summation modules coupled in series, each additional summation module receiving the outputs from a previous summation module, wherein the one or more additional summation modules includes a final summation module and wherein the final summation module outputs in parallel the correlation of the data word vector with each valid combination of the coefficient set; and a comparator coupled to the final summation module.
 40. The WLAN device of claim 39, wherein the one or more additional summation modules, include: a plurality of summers, each summer coupled to a different group of outputs from the previous summer, wherein each group of outputs from the previous summer represents a valid combination of outputs from the previous summation module.
 41. The WLAN device of claim 39, wherein the comparator includes: a plurality of magnitude comparators, wherein the plurality of magnitude comparators are arranged in a parallel tree configuration.
 42. The WLAN device of claim 39, wherein the comparator is a space splicing comparator.
 43. The WLAN device of claim 39, wherein the multiplier module includes a plurality of multipliers, one for each element of the data word vector.
 44. A wireless local area network (WLAN) device, comprising: a multiplier module having a coefficient set (C₀, . . . , C_(M-1)), wherein each coefficient in the coefficient set has one or more possible states, wherein the multiplier module receives a plurality of inputs, one for each element of an encoded data word vector (X₀, . . . , X_(M-1)), and multiplies each input with each state of a corresponding coefficient; a first summation module including a plurality of adders, wherein each adder receives a valid combination of multiplier module outputs and wherein an output of each adder in the first summation module includes two correlation components; a second summation module including a plurality of adders, wherein each adder receives a valid combination of first summation module outputs and wherein an output of each adder in the second summation module includes four correlation components; and a final summation module including a plurality adders, wherein each adder receives a valid combination of second summation module outputs and wherein the output of the final summation module is 64 valid correlations, each correlation having eight correlation components.
 45. The WLAN device of claim 44, wherein the first summation module has 16 adders, the second summation module has 32 adders, and the final summation module has 64 adders.
 46. The WLAN device of claim 44, wherein the device further comprises: a comparator coupled to the final summation module
 47. The WLAN device of claim 46, wherein the comparator includes: a plurality of magnitude comparators, wherein the plurality of magnitude comparators are arranged in a parallel tree configuration.
 48. The WLAN device of claim 46, wherein the comparator is a space splicing comparator.
 49. The WLAN device of claim 46, wherein the multiplier module includes a plurality of multipliers, one for each element of the data word vector.
 50. The WLAN device of claim 44, wherein the coefficient states have values of +1 and −1.
 51. The WLAN device of claim 44, wherein the coefficient states have values of +1, −1, +j, and −j. 